This invention relates to electronic instruments for detecting and/or measuring electrical signals and, more particularly, to electronic instruments for detecting and/or measuring the frequency spectrum of electrical signals. Specifically, one embodiment of the invention provides an integrated barium-ferrite tuned mixer particularly adaptable for use in an electronic instrument known as a spectrum analyzer. In one implementation, the mixer is preferably an even harmonic mixer which employs second harmonic mixing at frequencies from 26.5 to 31 GHz and fourth harmonic mixing at frequencies from 31 to 50 GHz. The mixer in accordance with one embodiment of the invention can also be combined with at least one additional barium-ferrite tuned resonator to form a barium-ferrite tuned resonator filter and mixer that allows for tuning over the frequency range of approximately 26.5 to 50 GHz.
Generally, a spectrum analyzer is a scanning receiver that displays power and modulation characteristics of electrical input signals over a selected frequency band. To cover a broad frequency range, e.g., from approximately 0 to 40 GHz, an input signal is typically divided into low-frequency and high-frequency portions. In this regard, the input signal is preferably effectively routed between low- and high-frequency signal processing sections of the spectrum analyzer depending upon the frequency of the input signal.
Accordingly, FIG. 1 shows a block diagram illustrating a superheterodyne receiver which forms the initial signal processing circuit of a conventional spectrum analyzer. Initially, all input signals, e.g., signals in the frequency range from approximately 0, e.g., 30 Hz, to 40 GHz, applied to an input 11, e.g., a coaxial connection, pass through a step attenuator 12 and are directed to a diplexer 13 which selectively routes the input signals to the low-frequency signal processing section on line 14 or to the high-frequency signal processing section on line 15.
On the one hand, low-frequency input signals, e.g., input signals having a frequency under 26.5 GHz, are applied to a low-frequency signal processing section of the spectrum analyzer. Input signals from approximately 0 to 26.5 GHz are diplexed to a front end 16, e.g., an HP 8563E spectrum analyzer available from Hewlett-Packard Company, Palo Alto, Calif., connected to a swept, yttrium-iron-garnet (YIG) tuned local oscillator (LO) 18 for spectrum analysis. Such a low-frequency signal processing section is disclosed in copending U.S. Patent application Ser. No. 08/094,833 entitled ROUTING YIG-TUNED MIXER filed on Jul. 20, 1993, in the name of Hassan Tanbakuchi and assigned to the same assignee as this patent application, the disclosure of which is hereby incorporated herein in its entirety by this reference.
On the other hand, high-frequency input signals, e.g., RF (millimeter) input signals having a frequency greater than 26.5 GHz, e.g., 26.5 to 40 GHz, are passed through a tunable bandpass filter 21. The passed RF input signal is then passed through a fixed attenuator 22 (e.g., a 6 dB attenuator) and downconverted by a harmonic mixer 23. The harmonic mixer 23 preferably combines the RF input signal with a signal 5produced by the LO 18, or a harmonic of the signal from the LO, to produce a predetermined IF output signal at a frequency, e.g., 321.4 MHz, suitable for further processing by the high-frequency analysis circuit (not shown) of the spectrum analyzer.
Considered in more detail, a spectrum analysis measurement is performed on a high-frequency input signal by sweeping the LO signal frequency over the frequency range of interest, while the predetermined IF frequency is monitored. The graph in FIG. 2 illustrates the result of the down conversion by the harmonic mixer 23 shown in FIG. 1, evidencing the relationship between the LO, RF, and predetermined IF frequencies. In FIG. 2, the vertical axis represents signal power, and the horizontal axis represents signal frequency. The predetermined IF signal 25 has a frequency equal to the difference between the LO signal (or harmonic) 27 and the RF input signal 29, so that the RF input signal is measured by monitoring a set IF frequency, below the LO signal frequency, at f.sub.RF =(n)f.sub.LO -f.sub.IF. However, an image RF signal above the LO signal frequency, at f.sub.RF =(n)f.sub.LO +f.sub.IF, will also produce a signal at the monitored IF frequency. To resolve this ambiguity, the tunable bandpass filter 21 shown in FIG. 1 acts as a tunable bandpass filter over a frequency range including f.sub.RF, as shown by the broken line curve 31 that appears in FIG. 2, thereby attenuating any image signal 33 at f'.sub.RF. Therefore, the passband of the tunable bandpass filter 21 shown in FIG. 1 must track the sweeping LO signal, with the center frequency of the passband separated from the LO frequency (or harmonic) by the IF signal frequency.
The tunable bandpass filter 21 shown in FIG. 1 is essential to eliminate unwanted mixing products from being displayed on the spectrum analyzer. Ferrimagnetic materials, such as YIG and barium-ferrite, are predominately used as magnetically tunable resonators for broadband (multi-decade) tunable filters. Filtering is accomplished by magnetically coupling RF signals to a spherical magnetic resonator. By placing the spherical magnetic resonator within the pole gap of an adjustable electromagnet, the tuned frequency of the resonator can be controlled. The ferrimagnetic resonance frequency for a spherical magnetic resonator is: EQU f.sub.r =.gamma. (H.sub.0 .+-.H.sub.a),
where H.sub.0 is the strength of the externally applied DC magnetic field produced by the electromagnet in ocrsteals, H.sub.a is the internal anisotropy field (in oersteds) within the ferrimagnetic material, and .gamma. is the gyromagnetic ratio (2.8 MHz/oersted).
The tunable bandpass filter 21 shown in FIG. 1 can be a YIG-tuned resonator filter, or preselector, which comprises a YIG sphere suspended between two orthogonal half loop conductors with the YIG sphere centered on the intersection of the loop axes. When the YIG sphere is not magnetized, an RF input signal is not transferred between the half loops, because there is no interaction between the RF input signal and the YIG sphere and the loops are perpendicular to each other. However, in the presence of an externally applied DC magnetic field produced by the electromagnet along an axis perpendicular to the half loops, the dipoles in the YIG sphere align with the DC magnetic field, producing a strong net magnetization, M, in the YIG sphere. An RF input signal applied to the input half loop therefore produces an alternating RF magnetic field perpendicular to the externally applied DC magnetic field, which causes the dipoles in the YIG sphere to precess around the DC magnetic field at the frequency of the RF input signal. The precession frequency is equal to the frequency of the RF input signal if the RF input signal frequency equals or closely approximates the dipole resonance frequency of the spherical YIG resonator.
That is, if an RF input signal at or near f.sub.r is applied to the input half loop, the YIG material exhibits ferrimagnetic resonance, such that the precessing dipoles create a circularly polarized magnetic field, rotating at the RF input signal frequency, in a plane perpendicular to the externally applied DC magnetic field. This rotating magnetic field couples to the output half loop, including an RF signal in the output half loop that, at the resonance frequency f.sub.r, is phase-shifted 90.degree. from the RF input signal.
The YIG-tuned resonator filter therefore acts as a gyrator. The phase shift in one direction through the YIG-tuned resonator filter differs from the phase shift in the other direction by 180.degree.. The filtering function is achieved because RF input signals deviating from the dipole resonance frequency by more than a small amount do not couple to the YIG sphere.
Because the resonance bandwidth can be made fairly narrow, the YIG resonator comprises a highly selective bandpass filter at RF frequencies, tunable by varying the strength of the externally applied DC magnetic field. Typical loaded Q values for YIG-tuned resonator filters range from 100 to 400.
On the one hand, YIG has traditionally been used as the ferrimagnetic material in tunable bandpass filters which operate up to 30 GHz. Moreover, YIG has been used as high as 40 GHz with some degradation in insertion loss and off-resonance isolation. There are, however, severe limitations in using YIG above 40 GHz. First, YIG has a relatively low saturation magnetization (i.e., &lt;1,750 Gauss). As a result, loop coupling is the only efficient topology to couple to a YIG sphere resonator at such high frequencies. However, designing a coupling loop with self-resonance frequencies above 40 GHz would require very small geometries, including sphere diameters less than 0.2 mm. Since the loop coupling coefficient is proportional to sphere volume and inversely proportional to the area enclosed by the coupling loop, smaller geometries decrease coupling coefficients (i.e., increase filter insertion loss) and decrease the filter bandwidth.
A second limitation in using YIG beyond 40 GHz is its small internal anisotropy field, H.sub.a, of about 100 to 200 oersteds. As the above equation demonstrates, a high magnetic field (e.g., &gt;10,000 oersteds) is required to tune YIG to frequencies beyond 30 GHz. This translates to problems with high power dissipation and magnetic saturation in the electromagnet.
On the other hand, barium-ferrite has been used effectively as the ferrimagnetic material in loop coupled tunable bandpass filters covering the frequency range of 26.5 to 40 GHz. Barium-ferrite has been successfully grown with an internal anisotropy field ranging from 7,500 to 15,000 oersteds depending upon doping. With a higher internal anisotropy field than YIG, barium-ferrite filters can be constructed with less stringent requirements on the electromagnet. For example, the same magnetic field needed to tune a YIG resonator to 28 GHz would tune a barium-ferrite resonator with H.sub.a =8,400 oersteds to 50 GHz, thereby covering the frequency range of 26.5 to 50 GHz. However, loop coupling to barium-ferrite is not suitable beyond 40 GHz due to loop self-resonance.
Also, barium-ferrite tunable bandpass filters are known operating from 26.5 to 75 GHz in waveguide bands. For example, waveguide-to-sphere plus iris coupling have been used to construct waveguide barium-ferrite tunable bandpass filters. See U.S. Pat. No. 4,888,569. While performance of such tunable bandpass filters has been proven, waveguide filters limit the filter operation to sub-octave bands (i.e., waveguide bandwidth). Also, waveguide structures are large and are therefore not suitable for incorporation into a portable spectrum analyzer. Furthermore, to achieve operation from 26.5 to 50 GHz, two switched waveguide filters and mixers are needed to cover respective frequency ranges of 26 to 40 GHz and 40 to 50 GHz.
The prior art circuit shown in FIG. 1 also has several additional drawbacks. RF input signals in the frequency range of 26.5 to 40 GHz are diplexed to the tunable bandpass filter 21 which is used as a preselector. A filtered signal is passed through the fixed attenuator 22 and applied to the harmonic mixer 23, where it is mixed with harmonics of the swept, YIG-tuned local oscillator (LO) 18 to produce the predetermined IF output signal. Since the tunable bandpass filter 21 and the harmonic mixer 23 are separated by coaxial cable, the fixed attenuator 22 is required to decrease the mismatch ripple between the filter and mixer. Therefore, the sensitivity of the spectrum analyzer is drastically reduced.
Additionally, a schematic diagram of one known waveguide harmonic mixer 23 is shown in FIG. 1A. This mixer comprises a tapered waveguide 24 to which the RF input signals are applied.
RF input signals at an input of the tapered waveguide 24 travel along the tapered waveguide to a reduced height output of the tapered waveguide, where an anti-parallel pair of diodes 25 is connected to one side of the tapered waveguide and a low-pass filter 26 comprising inductor L and capacitors C is connected to the other side of the tapered waveguide. An LO signal enters through a coaxial LO input, a capacitor 27, and the low-pass filter 26 to the anti-parallel pair of diodes 25. Mixing with the RF input signals occurs in the anti-parallel pair of diodes 25 as an even harmonic mixer. An IF signal produced in the anti-parallel pair of diodes 25 at 321.4 MHz is diplexed to an IF output port using the diplexing effect of an inductor 28 and the capacitor 27. This single-ended even harmonic mixer has the drawback of mixing broadband LO noise at 2f.sub.LO +f.sub.IF and 2f.sub.LO -f.sub.IF with the IF frequency, which increases the noise figure of the spectrum analyzer.